There are several different types or classes of audio amplifiers. A class A amplifier will conduct a DC current even though there is no audio signal present at the amplifier's input. The class A amplifier will also have a single polarity voltage swing. A class B amplifier is more efficient than a class A amplifier due to the fact that both of the class B amplifier's differential mode voltage swings are not on at the same time. Class AB amplifiers maintain a small bias current through complementary output drive transistors, so that the output voltage swing is centered about a ground voltage.
Class D amplifiers amplify the audio signal using digital processing techniques and drive a complementary output signal that is digital in nature, with an output voltage swing that is capable of moving fully from rail-to-rail where the output stage operates in switched mode. The output voltage waveform of a class D amplifier has a duty cycle that varies with the amplitude of the input signal.
With reference now to FIGS. 1a through 1c, there are shown diagrams illustrating a prior art modulator 100 with a pulse width modulated (PWM) output signal and prior art models of a non-feedback controlled PWM amplifier 150 and a feedback controlled PWM amplifier 175. The diagram shown in FIG. 1a illustrates a modulator 100 with a PWM output signal, commonly referred to as a PWM modulator. The PWM modulator 100 includes a comparator 105 that compares an input signal against a signal that is provided by a signal source 110 (the signal provided by the signal source 110 is commonly referred to as a carrier signal) and produces an output waveform that is a two-level PWM signal. The signal source 110 is typically configured to produce a carrier signal that is a triangular waveform. For example, the comparator 105 will produce an output signal with a +1 value (for example) if the input signal is greater than the carrier signal at the time of the comparison. If the input signal is less than the carrier signal when the comparison is made, then the comparator 105 will produce an output signal with a −1 value (for example). Since both the input signal and the carrier signal can be continually varying, the output signal of the comparator 105 can change between +1 and −1. The gain of the PWM modulator 100 can be modeled as the ratio between the peak-to-peak output of the PWM modulator 100 and the peak-to-peak amplitude of the output of the signal source 110 (the carrier signal).
Alternatively, the PWM modulator 100 can operate in a self-oscillating mode without the signal source 110 and the attendant carrier signal. This can occur by applying a positive feedback signal around the comparator 105 (i.e., adding hysteresis) and/or by establishing a suitable feedback path around the comparator 105 with a combination delay and phase delay that can cause the PWM modulator 100 to oscillate and produce a two-level square wave signal out from the comparator 105.
The diagram shown FIG. 1b illustrates a prior art non-feedback controlled PWM amplifier 150. The non-feedback controlled PWM amplifier 150 includes the PWM modulator 100 along with a switching power stage 155, which typically comprises high power switches (such as MOSFETs) connected to a power supply (not shown). The switching power stage 155 produces a high power PWM signal that can drive a load 160 (a speaker, for example). With the use of the high power switches, a class D amplifier can achieve very high power efficiency, which is a significant advantage of class D amplifiers over traditional class A and class AB amplifiers.
Additionally, an inductor-capacitor (LC) filter 165 can be inserted between the switching power stage 155 and the load 160. The LC filter 165 can be used to filter out the high frequency components of the PWM signal that may otherwise cause undesired effects (such as, heating the speaker voice coil and radiating electromagnetic interference (EMI)). An ideal LC filter 165 will not dissipate energy but will only block the passage of high frequency signals. A disadvantage of the LC filter 165 is that it has a frequency response that is a function of the load impedance, i.e., the LC filter 165 has a high output impedance that tends towards infinity at the resonant frequency of the LC filter 165.
In practice, the non-feedback PWM amplifier 150 can be susceptible to non-ideal electrical behavior, such as non-harmonic components, noise and distortion arising from switching delays, non-linearities, distortions in the carrier signal, amplitude distortion, power supply noise and ripple, and so forth. The non-ideal electrical behavior can cause the PWM modulator 100 to introduce noise and distortion into the output signal. A feedback control loop added to the non-feedback PWM amplifier 150 can provide compensation for many of the problems introduced by the non-ideal electrical behavior and produce an amplifier with better performance characteristics.
The diagram shown in FIG. 1c illustrates a prior art feedback controlled PWM amplifier 175. The feedback controlled PWM amplifier 175 includes the components of the non-feedback controlled PWM amplifier 150, including the PWM modulator 100 with power switching stage 155 and the LC filter 165 and can be coupled to the load 160. The feedback controlled PWM amplifier 175 also includes a loop filter 180 and multiple feedback loops, including a first feedback loop 182 from the output of the power switching stage 155, a second feedback loop 184 from the LC filter 165, and a third feedback loop 186 from the output of the LC filter 165. The feedback loops bring the signals to the loop filter 180, where the loop filter 180 implements a summing function that subtracts the feedback signals from the input signal and applies various filter transfer functions to shape the feedback controlled PWM amplifier's frequency response to control feedback suppression and system (the overall feedback controlled PWM amplifier 175) stability.
The loop filter 180 may use feedback signals from both the output of the LC filter 165 (the third feedback loop 186) and the input of the LC filter 165 (the first feedback loop 182) as well as the feedback signals derived from the LC filter 165 itself (the second feedback loop 184). The feedback signals derived from the LC filter 165 can typically be inductor or capacitor currents. Generally, the more feedback taken from the output of the LC filter 165 (the third feedback loop 186), the lower the closed loop output impedance of the feedback controlled PWM amplifier 175. However, the LC filter 165 can add a large phase delay of up to 180 degrees which can make it difficult to achieve a stable loop when the feedback is primarily taken from after the LC 165.
The feedback controlled PWM amplifier 175 can be characterized by its loop transfer function, H(s). The loop transfer function, H(s), can be obtained by conceptually opening the loop (at the output of the PWM modulator 100, for example) and calculating the transfer function of a path along the loop, around to the output of the PWM modulator 100. Therefore, in reference to FIG. 1c, the loop transfer function is a function of the loop filter 180, the PWM modulator 105, and the power switching stage 155. If loop transfer functions of feedback paths 184 and 186 were computed, the loop transfer functions would include the LC filter 165 and the load 160, respectively.
The loop transfer function provides a measure of the stability of the system (the feedback controlled PWM amplifier 175) and its ability to suppress errors, including those in the PWM modulator 105 and the power switching stage 100. The loop transfer function and its derivation is considered to be well understood by those of ordinary skill in the art of the present invention and will not be further discussed herein.
One disadvantage of the prior art is that the feedback loop takes its signal primarily before the LC filter 165, so the output impedance of the feedback controlled PWM amplifier 175 at high frequencies can be high. In a typical LC filter 165, the output impedance is highest at the frequency band around the resonant frequency of the LC filter 165. The resonant frequency of the LC filter 165 is usually placed well above the audible range; therefore, the output impedance is highest at the upper ranges of the audio frequency range. The variation in the output impedance can alter the frequency response of the LC filter 165, and therefore the frequency response of the feedback controlled PWM amplifier 175 when different loads are applied to the feedback controlled PWM amplifier 175.
Another disadvantage of the prior art is that a typical loop filter 180 will have active components, such as operational amplifiers, which could have slew-rate distortion. Therefore, a limit may need to be placed on the switching frequency of the feedback controlled PWM amplifier 175 in order to reduce slew-rate distortion, which would require the use of expensive active circuits.
Yet another disadvantage of the prior art is that aliasing error may not be minimized, resulting in higher than necessary distortion levels. The aliasing error can increase the overall distortion of the feedback controlled PWM amplifier 175.
A further disadvantage of the prior art is that feedback loops that include the LC filter 165 in the loop can be unstable when the feedback controlled PWM amplifier 175 is not loaded. A prior art technique to solve this problem is to place a so-called Zobel network on the output terminal of the feedback control PWM amplifier 175. The Zobel network provides resistive loading near the resonant frequency of the LC filter 165. The Zobel network can be a serial connection of a resistor and a capacitor with a time constant that is compatible with the time constant of the LC filter 165. However, the Zobel network will dissipate a substantial amount of power when a large amplitude audio signal is being reproduced. This reduces the overall power efficiency of the feedback controlled PWM amplifier 175 and requires bulky components as well as protection schemes to prevent damage to the components of the Zobel network due to overheating.